1. Field of the Invention
The present invention relates to an automatic frequency control method and an apparatus therefor, and more particularly, to an automatic frequency control technology which is used for detecting sync signals within a signal which is modulated and transmitted according to M-ary phase-shift-keying (MPSK).
2. Description of the Related Art
In the field of communication technology, a receiver employing a sync detection method detects a sync signal within a received signal and then demodulates the signal. It is possible to perfectly detect the sync signal and to perfectly demodulate the received signal into an original signal when both the frequency and the phase of a local carrier signal are the same as those of a modulated carrier signal. Thus, both the frequency and the phase of a local carrier signal should be synchronized, respectively, with those of a modulated carrier signal. A conventional technology of detecting the sync from the received signal will be described below with reference to FIGS. 1, 2A and 2B.
FIG. 1 shows an automatic frequency control apparatus for general quadrature phase-shift-keying (QPSK) demodulation. In FIG. 1, a received signal is input to multipliers 11 and 12. Each multiplier 11 and 12 multiplies the received signal by an output signal from a voltage controlled oscillator 19. Multipliers 11 and 12 generate signals having phases different from each other by 90.degree.. The outputs of multipliers 11 and 12 are low-frequency-converted into baseband signals by low-pass filters 13 and 14, respectively. When the received signal does not include Gaussian noise, the received signal can be expressed by the following equation: S(t)=A cos .omega..sub.0 t+.phi.(t). Here, S(t) is a signal which is a function of time t, where A is an amplitude, .omega..sub.0 is a frequency, and .phi.(t) is a phase which is also a function of time t. In this case, low-pass filter 13 generates an in-phase (I) channel signal which is represented by the equation: I(t)=A cos (.DELTA..omega.t+.phi.(t)).
On the other hand, low-pass filter 14 generates a quadrature (Q) channel signal which is represented by the equation: Q(t)=A sin (.DELTA..omega.t+.phi.(t)). Here, A is an amplitude, .DELTA..omega. is a frequency difference, and .phi.(t) is a phase which is a function of time t. Signals I(t) and Q(t) after passing through low-pass filters 13 and 14 are digitally converted by analog-to-digital converters 15 and 16, respectively. The output signals I.sub.k and Q.sub.k of analog-to-digital converters 15 and 16 are transmitted for signal demodulation and simultaneously supplied to a frequency detector 17. The output signals I.sub.k and Q.sub.k of analog-to-digital converters 15 and 16 have a phase .phi.(t) which is varied for every symbol period T.sub.b. For example, in case of the QPSK method, phase .phi.(t) has a value of one of 45.degree., 135.degree., -45.degree. and -135.degree., in which .phi.(t) is varied for each symbol period T.sub.b according to bit-stream information from a transmitter end.
Frequency detector 17 receives two channel signals I.sub.k and Q.sub.k and generates a frequency offset signal V(k). Frequency offset signal V(k) is generated when the frequency of a local oscillator does not match the frequency of the received signal in the actual apparatus. Frequency offset signal V(k) passes through a loop filter 18, and then is supplied to a voltage controlled oscillator (VCO) 19. VCO 19 generates local oscillation signals having different frequencies, based on the input frequency offset signal V(k). The local oscillation signals are supplied to multipliers 11 and 12, and are used for generating an I-channel signal and a Q-channel signal, each signal having a 90.degree. phase difference from the other. By repeating such a process, both the frequency and the phase of the internally generated oscillation signals are synchronized with those of the received signal.
The technologies of detecting the frequency offset information are described in the following references:
[1] AFC Tracking Algorithms (IEEE Trans. on Communications, Vol. COM-32, No. 8, August 1984, pp. 935-947); and PA1 [2] A New QPSK Demodulator For Digital DBS Receivers (IEEE 1922, pp. 192-193). PA1 sampling the received signal with a predetermined sampling frequency and generating a sampled complex signal; PA1 detecting a phase difference value between the sampled complex signal and a previously received sampled complex signal, to generate a first phase difference detection signal having a phase value corresponding to the detected phase difference value; PA1 altering the detected phase difference value of the first phase difference detection signal by a phase altering factor, to generate a second phase difference detection signal having a phase value corresponding to the altered phase difference value; PA1 determining transmission phase information of the received signal by using the altered phase difference value of the second phase difference detection signal and reference phase values used for information transmission of the M-ary phase-shift-keying modulation signal; and PA1 generating the frequency offset signal by using the determined transmission phase information and the altered phase difference value. PA1 a sampler for sampling the received signal with a predetermined sampling frequency and generating a currently sampled complex signal; PA1 a phase difference detector for receiving the currently sampled complex signal and detecting a phase difference value between the currently sampled complex signal and a previously received sampled complex signal, to generate a first phase difference detection signal having a phase value equal to the detected phase difference value; PA1 a phase difference altering unit for altering the detected phase difference value of the first phase difference detection signal by a phase altering factor, and generating a second phase difference detection signal having a phase value equal to the altered phase difference value; PA1 a circuit for determining transmission phase information from the altered phase difference value by using the altered phase difference value of the second phase difference detection signal and reference phase values used for information transmission in the M-ary phase-shift-keying modulation; and PA1 a generator for generating the frequency offset signal based on the determined transmission phase information and the altered phase difference value.
FIG. 2A shows a circuit which employs a cross-product means disclosed in the above reference [1] as the frequency detector 17 of FIG. 1. When the sampled and digitally converted I-channel and Q-channel signals I.sub.k and Q.sub.k are input to the circuit shown in FIG. 2, I-channel signal I.sub.k is supplied to a delay 21 and multiplier 24, while Q-channel signal Q.sub.k is supplied to a delay 22 and multiplier 23. Multiplier 23 multiplies the delayed I-channel signal I.sub.k-1 by Q-channel signal Q.sub.k, while multiplier 24 multiplies the delayed Q-channel signal Q.sub.k-1 by I-channel signal I.sub.k. A subtractor 25 subtracts output signal I.sub.k.Q.sub.k-1 of multiplier 24 from output signal Q.sub.k.I.sub.k-1 of multiplier 23. Subtractor 25 generates a frequency offset signal V(k) which is determined by a sampling period T.sub.s, where frequency offset signal V(k) is generated according to the following equation (1). EQU V(k)=A.sup.2 sin (.DELTA..omega.T+.theta..sub.k) (1)
Here, .DELTA..omega.=.omega..sub.1 -.omega..sub.0, .phi..sub.(t) =.phi..sub.k, kT.sub.s .ltoreq.t.ltoreq.(k-1)T.sub.s, and .theta..sub.k =.phi..sub.k -.phi..sub.k-1. When T.sub.s &lt;T.sub.b (where T.sub.b =nT.sub.s), that is, when the input signal is oversampled, frequency offset signal V(k) generated by subtractor 25 is expressed by the following equation (2). EQU V(k)=A.sup.2 sin (.DELTA..omega.T.sub.s), if k.noteq.nl (l is an integer), and EQU V(k)=A.sup.2 sin (.DELTA..omega.T.sub.s +.theta..sub.1), if k=nl (l is an integer) (2)
FIG. 2B shows a frequency detector using an arc-tangent means which is disclosed in reference [2]. Arc-tangent means 27 receives two channel signals I.sub.k and Q.sub.k and performs an arc-tangent operation using Q-channel signal Q.sub.k as a numerator and I-channel signal I.sub.k as a denominator. A differentiator 28 generates a frequency offset signal V(k) which is expressed by the following equations (3) and (4), based on the output signal from arc-tangent means 27. EQU V(k)=.DELTA..omega.T.sub.s +.theta..sub.k ( 3) EQU V(k)=.DELTA..omega.T.sub.s, if k.noteq.nl (l is an integer), and EQU V(k)=.DELTA..omega.T.sub.s +.theta..sub.1, if k=nl (l is an integer)(4)
The above equation (3) represents an output signal of differentiator 28 when T.sub.s =T.sub.b. The above equation (4) represents an output signal of differentiator 28 when T.sub.s &lt;T.sub.b (where T.sub.b =nT.sub.s), that is, when the signal is oversampled. To detect an exact frequency, it is not desirable to include terms of .theta..sub.k and .theta..sub.1 of which the values are varied according to the transmitted information. However, as it can be seen from the above equations (1) and (3), when the sampling frequency equals the symbol rate, frequency offset signal V(k) includes a transmission phase value .theta..sub.k. Therefore, it becomes impossible to detect the exact frequency which is proportional to only frequency offset information .DELTA..omega.T. Also, when the oversampled sample has a symbol transition, transmission phase value .theta..sub.k exists in frequency offset signal V(k), which interferes with the exact frequency detection. Although the performance of the frequency detection can be enhanced according to a degree of oversampling thereof, such oversampling causes an increased cost of hardware for oversampling when the symbol rate is more than 20 MHz as in a direct broadcasting satellite (DBS). Such a problem occurs in a differentiator automatic frequency control apparatus, as well as in a discrete Fourier transform automatic frequency control apparatus.